The invention relates generally to acoustic resonators and more particularly to controlling the effective coupling coefficient of a film bulk acoustic resonator.
In many different communications applications, a common signal path is coupled to both an input of a receiver and an output of a transmitter. For example, in a cellular or cordless telephone, an antenna may be coupled to the receiver and the transmitter. In such an arrangement, a duplexer is often used to couple the common signal path to the input and the output. The function of the duplexer is to provide the necessary coupling to and from the common signal path, while preventing the signals generated by the transmitter from being coupled to the input of the receiver.
One type of duplexer is referred to as the half duplexer. A half duplexer uses a switch to connect the common signal path to the receiver or the transmitter on a time division basis. The half duplexer has the desired coupling and attenuation properties, but is unacceptable in many telephony applications, since it does not allow parties of a call to speak and be heard simultaneously.
A type of duplexer that is more acceptable for telephony applications is the full duplexer. A full duplexer operates only if the transmit signal has a frequency that is different than the frequency of the receive signal. The full duplexer incorporates band-pass filters that isolate the transmit signal from the receive signal according to the frequencies. FIG. 1 illustrates a conventional circuit used in cellular telephones, personal communication system (PCS) devices and other transmit/receive devices. A power amplifier 10 of a transmitter is connected to a transmit port 12 of a full duplexer 14. The duplexer also includes a receive port 16 that is connected to a low noise amplifier (LNA) 18 of a receiver. In addition to the transmit port and the receive port, the duplexer 14 includes an antenna port 20, which is connected to an antenna 22.
The duplexer 14 is a three-port device having the transmit port 12, the receive port 16 and the antenna port 20. Internally, the duplexer includes a transmit band-pass filter 24, a receiver band-pass filter 26 and a phase shifter 28. The passbands of the two filters 24 and 26 are respectively centered on the frequency range of the transmit signal that is input via the power amplifier 10 and the receive signal to which the receiver is tuned.
The requirements for the band-pass filters 24 and 26 of the duplexer 14 are stringent. The band-pass filters must isolate low intensity receive signals generated at the antenna 22 and directed to the input of the low noise amplifier 18 from the strong transmit signals generated by the power amplifier 10. In a typical embodiment, the sensitivity of the low noise amplifier may be in the order of xe2x88x92100 dBm, while the power amplifier may provide transmit signals having an intensity of approximately 28 dBm. It is expected that the duplexer 14 must attenuate the transmit signal by approximately 50 dB between the antenna port 20 and the receive port 16 to prevent any residual transmit signal mixed with the receive signal at the receive port from overloading the low noise amplifier.
One type of PCS that is used in a mobile telephone employs code division multiple access (CDMA). The CDMA PCS wireless bands are centered at approximately 1920 MHz and have an especially stringent regulatory requirement for duplexer performance. Some concerns will be identified with reference to FIG. 2. A passband 30 is defined by several poles and several zeros. The poles and zeros are equidistantly spaced from a center frequency 32. For the transmitter passband 30, the transmitter-to-antenna insertion loss 34 is preferably better than xe2x88x923 dB over most of the band. The isolation from the transmitter to receiver ports exceeds 50 dB across most of the transmitter band and 46 dB in the receiver band. The crossover between the transmitter and receiver bands occurs around 1920 MHz. The transmitter and receiver bands are approximately 3.0 percent of the carrier frequency, so that extremely sharp filter roll-off 36 and 38 is required. As will be explained more fully below, the lower-frequency poles and zeroes and the roll-off 36 are determined by the characteristics of shunt resonators, while the higher-frequency poles and zeroes and the roll-off 38 are determined by the characteristics of series resonators.
Another challenge for the duplexer is achieving power handling requirements. The power amplifier 10 of FIG. 1 in the transmitter can deliver 1 Watt of power to the transmit port 12 of the duplexer 14. The band-pass filter 24 must be capable of handling such power without being destroyed and without its performance being degraded.
The duplexer 14 will be described in greater detail with reference to FIG. 3. The duplexer includes a transmit film bulk acoustic resonator (FBAR) array 40 and a receive FBAR array 42. The transmit FBAR array is a two-stage ladder circuit having two series FBARs 44 and 46 and two shunt FBARs 50 and 52. The series FBARs are connected in series between the transmit port 12 and the antenna port 20, while the shunt FBARs are connected between electrical ground and nodes between the series FBARs. Each full stage of an FBAR array is composed of one series FBAR and one shunt FBAR. In order to handle the high power generated by the power amplifier at the filter input of the transmit filter, power bars are used for each of the series elements 44 and 46.
The receive FBAR array is a 3xc2xd-stage ladder circuit. A half stage is limited to either one series FBAR or one shunt FBAR. In the exemplary array 42, the half stage is a shunt FBAR 60. The FBAR array includes three series FBARs 54, 56 and 58 and four shunt FBARs 60, 62, 64 and 66. The series FBARs are connected in series between the ninety degree phase shifter 28 and the receive port 16. The shunt FBARs are connected between electrical ground and nodes between the series FBARs.
Circuits suitable for use as the ninety degree phase shifter 28 are known in the art. As examples, the phase shifter may be composed of inductors and capacitors or may be a xcex/4 transmission line.
Within the transmit FBAR array 40, each series FBAR 44 and 46 may have the same resonant frequency (frTx), which may be centered at 1880 MHz. Similarly, the shunt FBARs 50 and 52 may have the same resonant frequency, but the resonant frequency of the series FBARs is approximately 1.0 percent to 3.0 percent (typically 1.6 percent) greater than that of the shunt FBARs. As a result, the poles that were described with reference to FIG. 2 are provided.
The receive FBAR array 42 of the receive band-pass filter 26 may also be composed of series FBARs 54, 56 and 58 having the same frRx and shunt FBARs 60, 62, 64 and 66 having the same resonant frequency that is 3.0 percent different than the resonant frequency frRx of the series FBARs. Here, frRz is centered at 1960 MHz.
Other considerations that affect the shape of the response shown in FIG. 2 are the figure of merit, which is referred to as Q, and the effective coupling coefficient, which is also referred to as kt2. The effective coupling coefficient may be considered as being the ratio of electrical energy to acoustic energy in the operation of a particular FBAR. It has been the goal to maximize both Q and the effective coupling coefficient. As a result of the fabrication process, the effective coupling coefficient can be as high as 8.0 percent. It has been experimentally determined that Q is dependent upon kt2 and, in some cases, that it is better to decrease kt2 in order to significantly increase Q. The Q determines the roll-off of the response.
What is needed is a fabrication method and a resulting duplexer which provide a very steep roll-off in the operation of an array of acoustic resonators.
The performance of arrays of acoustic resonators is enhanced by tailoring the effective coupling coefficients of the individual acoustic resonators on the basis of the functions of the resonators. In a duplexer embodiment, the effective coupling coefficients of FBARs in a transmit band-pass filter are fabricated to have a lower effective coupling coefficient than the FBARs of the receive band-pass filter of the same duplexer.
In one embodiment, the difference in the effective coupling coefficients is achieved by varying the thicknesses of the electrode layers. For a given frequency, the effective coupling coefficient of an acoustic resonator is modified by varying the ratio of the thickness of the piezoelectric layer to the total thickness of the electrode layers. Typically, a goal in the fabrication of FBARs is to minimize the thickness of the electrode layers, thereby providing an xe2x80x9cintrinsicxe2x80x9d effective coupling coefficient. For example, this intrinsic coefficient may be in the range of 7.0 percent to 8.0 percent. However, the coupling coefficient of an FBAR filter having a given resonant frequency can be adjusted downwardly by decreasing the ratio of the thickness of the piezoelectric layer to the total thickness of the electrode layers, since the resonant frequency is dependent upon the xe2x80x9cweighted thicknessxe2x80x9d (i.e., the physical thickness weighted on the basis of the selection of electrode and piezoelectric materials) of the electrode-piezoelectric stack. As one example of a transmit filter, the thickness of molybdenum (Mo) electrodes can be increased and the thickness of aluminum nitride (AIN) can be reduced in order to achieve a degraded effective coupling coefficient in the range of 2.5 percent to 4.0 percent, while maintaining a targeted resonant frequency. Similarly, a receive filter can be fabricated to have an effective coupling coefficient in the range of 4.0 percent to 6.0 percent by selecting the appropriate thicknesses for the layers that form the FBARs of the receive filter.
The method of fabricating an array of acoustic resonators in accordance with this embodiment includes a step of selecting a first target frequency range and a first target effective coupling coefficient for operation of an FBAR transmit (Tx) filter, and includes selecting a second target frequency and a second target coupling coefficient for operation of an FBAR receiver (Rx) filter. The thicknesses and materials of the piezoelectric and electrode layers for forming the two FBAR filters are determined on the basis of achieving the target resonant frequencies and the target effective coupling coefficients. The determinations include selecting an increased electrode layer thickness for at least one electrode of the Tx FBARs, so that the Tx FBAR filter will have the degraded coefficient. The two filters are then formed according to the selected thicknesses and materials.
In the fabrication of the two filters, the electrode material may be Mo and the piezoelectric material may be AIN. Using these materials, the electrode layers of the FBAR Tx filter having the degraded coupling coefficient will have a thickness that can be in the range of 1.2 to 2.8 times the thickness of the electrode layers of the Rx filter with the higher coefficient. For example, in a communications device that is compatible with the CDMA PCS standard, the Rx filter may have electrode layer thicknesses of 2200 xc3x85 and a piezoelectric thickness of 2.2 microns in order to achieve a coupling coefficient in the range of 5.6 percent to 5.8 percent, while the Tx filter may have electrode layer thicknesses of 4500 xc3x85 and a piezoelectric thickness of roughly 8000 xc3x85 in order to achieve a coupling coefficient in the range of 3.1 percent to 3.2 percent. The Q (and therefore the steepness of the roll-off) is almost two times higher for the Tx filter than for the Rx filter.
In one application, a desired filter arrangement of FBARs is designed to include at least one xe2x80x9cpower barxe2x80x9d in order to increase the power handling capacity along a path of the filter arrangement. A xe2x80x9cpower barxe2x80x9d is defined herein as a pair of large area FBARs which are connected in series in place of a single target FBAR. Each large area FBAR occupies an area that is twice the area of the target FBAR. The parallel-series combination defined by the power bar (in the series connection of conventional electrical equivalent circuits) allows the impedance of the power bar to remain at the target impedance of the target FBAR, but reduces the power density by a factor of four.
In a second embodiment of the invention, the degraded effective coupling coefficient is achieved by forming a capacitor in parallel with at least some of the resonators of the Tx filter. Preferably, the capacitor is formed using materials that are deposited in steps for fabricating the array of acoustic resonators. For example, the electrodes and the piezoelectric layer that are deposited to fabricate the FBARs may be utilized in the formation of a capacitor that is placed in parallel with at least one FBAR of the Tx filter to degrade the effective coupling coefficient. However, the concern in using these layers is that a resonator will be fabricated, rather than a capacitor. One method for ensuring that the additional component functions as a capacitor is to fabricate the electrode-piezoelectric stack of the component directly on the substrate, rather than suspending the stack. In this manner, the substrate provides the means for mass loading the capacitor, thereby pulling the frequency off center.
A second method is to use the gold layer, which is conventionally used to provide contact pads, as the means to pull the resonator component off frequency. This second method is preferred, since the first method may form a high loss capacitor, while the second method is the one that will form a high Q component. By utilizing the gold layer and by suspending the capacitor component as a free-standing membrane in the same manner as the FBARs, the capacitor functions as a high Q resonator, but at a much lower frequency than the first and second FBARs. An advantage is that the frequency of the capacitor can be xe2x80x9ctunedxe2x80x9d to not only be displaced from the frequency of interest, but to form a parasitic resonance at frequencies where the duplexer does not perform well. As one example, the capacitor may resonate at 1510 MHz, which is a frequency at which existing duplexers do not perform well in the rejection of energy. Tuning the capacitor to 1510 MHz allows a designer to incorporate specific shunt and series type resonators that reduce leakage of the 1510 MHz signal. This is achieved without any additional process steps to the FBAR fabrication. The tuning of the capacitor can be provided merely by properly selecting the thickness of the gold and other layers in the electrode-piezoelectric stack of the capacitor.
An advantage of the methods described above is that the performance of an array of acoustic resonators is enhanced without significantly affecting the fabrication process. By tailoring the effective coupling coefficients of individual resonators within a full duplexer, roll-off at the opposite edges of the passband can be tailored.